Fast locking mechanism for channelized ultrawide-band communications

ABSTRACT

A receiver for acquisition and lock of an impulse radio signal comprising an adjustable time base to output a sliding periodic timing signal having an adjustable repetition rate, and a decode timing modulator to output a decode signal in response to the periodic timing signal. The impulse radio signal is cross correlated with the decode signal to output a baseband signal. The receiver integrates T samples of the baseband signal and a threshold detector uses the integration results to detect channel coincidence. A receiver controller stops sliding the time base when channel coincidence is detected. A counter and extra count logic, coupled to the controller, are configured to increment or decrement the address counter by a one or more extra counts after each T pulses is reached in order to shift the PN code modulo for proper phase alignment of the periodic timing signal and the received impulse radio signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.10/411,090, filed Apr. 11, 2003, which is a continuation of U.S. patentapplication Ser. No. 09/158,570, filed Sep. 22, 1998, which is acontinuation of U.S. patent application Ser. No. 08/761,602, filed Dec.6, 1996, now U.S. Pat. No. 5,832,035, entitled “Fast Locking Mechanismfor Channelized Ultrawide-Band Communication,” which is an FWC of U.S.patent application Ser. No. 08/487,990, filed Jun. 7, 1995, nowabandoned, which is a continuation-in-part of commonly owned, co-pendingU.S. patent application Ser. No. 08/309,973, filed Sep. 20, 1994, nowU.S. Pat. No. 5,677,927, entitled “Ultrawide-Band Communications Systemand Method,” and U.S. patent application Ser. No. 08/428,489, filed Apr.27, 1995, now U.S. Pat. No. 5,687,169, entitled “Full DuplexUltrawide-Band Communications System and Method,” all of which areincorporated herein by reference and to which 35 U.S.C. §120 priority ishereby claimed.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to the field of communications, and moreparticularly, the present invention relates to a fast locking mechanismfor channelized ultrawide-band communications.

2. Related Art

Conventional transceivers operating with narrow band signals typicallyuse the same antenna to transmit and receive signals. The transmit andreceive signals are usually the same or very close in frequency.Switching between the transmit and receive mode can be done at very highrates, depending on the density of each packet of data.

Full duplex operation has traditionally been accomplished by eitherfrequency domain or a time domain multiple access (FDMA or TDMA). Inorder to isolate the transmitter and receiver, FDMA uses frequencyfilters and hybrids, while TDMA uses a duty cycle scheme in which thetransmitter and receiver alternate operation.

An example of an FDMA full duplex voice communication system is anamateur radio transceiver that operates with different transmit andreceive frequencies. For example, the separated frequencies could be 144Mhz and 436 Mhz. In such a system, the antennas are usually different,and filters must be used in the receiver to eliminate transmitter noisefrom the adjacent transmit antenna. Otherwise, the receiver could easilybe overloaded by its own transmitter.

Impulse radio technology, on the other hand, is ultrawide-band bydefinition. The original descriptions of impulse radio may be found in anumber of United States Patents by the present inventor. Three of theseare U.S. Pat. Nos. 4,641,317 (issued Feb. 3, 1987), 4,813,057 (issuedMar. 14, 1989) and 4,979,186 (issued Dec. 18, 1990). Because of theultrawide-band characteristics of impulse radio, it is difficult tomodify impulse radio systems to use conventional duplex schemes.

In order to achieve full duplex in impulse radio technology, separatetransmit and receive antennas are required for hand-held transceiverapplications. This is because the receiver can not be disconnected fromthe antenna fast enough to permit transmission using the same antenna.Therefore, the size of the impulse radio antennas must be relativelysmall.

An impulse radio system with many users communicating with one anotherrequires that they all have the same size antennas. In addition, forimpulse radio communications in the same bandwidth, it is assumed thatthe transmit and receive antennas are the same size as well. Theseconstraints complicate the implementation of full duplex in impulseradio technology, because both the transmitter and receiver are usuallyoperated in the same ultrawide frequency bandwidth.

Impulse radio technology permits operation at rates so high that thereis no time for the signal to reach the intended receiver before the nextpulse is transmitted. This situation causes several pulses to be presentin the space between the two transceiver units. When there is motionbetween them such as in mobile communications, there occurs theunavoidable condition wherein the transmitter and receiver must operatesimultaneously.

In order to operate in full duplex mode in a mobile environment, thetransmitter and receiver would be required to operate simultaneouslywhenever the distance separating them increases or decreases by amultiple of C/R, where C is the speed of light and R is the repetitionrate. For example, if R=1 million pulses per second, these zones wouldbe about 300 meters, and so on. Although full duplex mode of operationis very desirable, this effect makes it unpractical to do so.

In order for pairs of users to simultaneously communicate independently,some form of channelization is required to avoid cross-talk. Onechannelization technique is to use different pulse repetition rates foreach pair of transceivers that communicate in proximity of othertransceivers. This technique, however, has limited channel capacity aslimited discrete pulse repetition rates are actually available forimpulse radio communications and may interfere with other communicationservices.

A second approach to channelization is to use different pseudo randomnoise (PN) codes. According to this technique, the number of channelsfor impulse radio communications is only limited by the complexity anduniqueness of orthogonal (i.e., non-interfering) PN codes. The inherentcomplexity of using PN codes for channelization is that the codes mustbe identifiable (i.e., acquisitioned and locked) and decoded in a shortperiod of time for full duplex communications to be realized.

What is needed for this PN coded approach is an acquisition mechanismthat is applicable to impulse radio technology, and that permits fastlocking of impulse radio signals.

SUMMARY OF THE INVENTION

The present invention is directed to a fast locking mechanism forchannelized ultrawide-band communications in an impulse radio receiver.An acquisition and lock method includes sliding a periodic timingsignal. A decode signal is produced using the periodic timing signal,wherein successive decode signals are coded by successive chips of apseudo noise (PN) code having a predetermined modulo length. A receivedimpulse radio signal is cross correlated with the decode signal tooutput a baseband signal. T samples of the baseband signal areintegrated to output an integration result that is then compared with athreshold value to output a channel coincidence signal.

If channel coincidence (i.e., acquisition) is detected, a constantrate-control signal is output to stop the periodic timing signal fromsliding. Otherwise, the periodic timing signal is adjusted, andsuccessive trials of T pulses of the periodic timing signal areintegrated and threshold detected until channel coincidence is detected.Typically, the process is stop acquisition if the entire PN code modulolength is completed before channel coincidence is detected.

A receiver for acquisition and lock of an impulse radio signal comprisesan adjustable time base to output the periodic timing signal having anadjustable repetition rate, and a decode timing modulator to output thedecode signal in response to the periodic timing signal. A crosscorrelator in the receiver cross correlates the impulse radio signalwith the decode signal to output a baseband signal.

The receiver integrates T samples of the baseband signal and a thresholddetector uses the integration results to detect channel coincidence. Areceiver controller stops sliding the time base when channel coincidenceis detected. A counter and extra count logic, coupled to the controller,are configured to increment or decrement the address counter by a one oror more extra counts. This count adjustment is made after each T pulsesis reached in order to shift the PN code modulo for proper phasealignment of the periodic timing signal and the received impulse radiosignal.

In an alternative embodiment, plural decode signals are generated andare cross correlated with received impulse radio signals using aplurality of cross correlators to reduce the time to acquire channellock. In still a further embodiment, a fast cross correlator can beused.

BRIEF DESCRIPTION OF THE FIGURES

FIGS. 1A and 1B show a 2 GHz center frequency monocycle pulse in thetime and frequency domains, respectively, in accordance with the presentinvention.

FIGS. 2A and 2B are illustrations of a 1 mpps system with 1 ns pulses inthe time and frequency domains, respectively, in accordance with thepresent invention.

FIG. 3 illustrates a modulating signal that changes the pulse repetitioninterval (PRI) in proportion to the modulation in accordance with thepresent invention.

FIG. 4 is a plot illustrating the impact of pseudo-random dither onenergy distribution in the frequency domain in accordance with thepresent invention.

FIG. 5 illustrates the result of a narrowband sinusoidal (interference)signal overlaying an impulse radio signal in accordance with the presentinvention.

FIG. 6 shows the “cross correlator” transfer function of an impulseradio receiver in accordance with the present invention.

FIG. 7 illustrates impulse radio multipath effects in accordance withthe present invention.

FIG. 8 illustrates the phase of the multipath pulse in accordance withthe present invention.

FIG. 9 shows a representative block diagram of a full duplex impulseradio system, in accordance with the present invention.

FIG. 10 shows timing of transmitted and received pulses at atransceiver.

FIG. 11 shows contention zones between an impulse radio transmitter andreceiver.

FIG. 12 shows a delay transmit technique to minimize the affect ofcontention zones between an impulse radio transmitter and receiver, inaccordance with an embodiment of the present invention.

FIG. 13 shows a flow diagram for a pulse interleaving technique for fullduplex impulse radio communications, in accordance with an embodiment ofthe present invention.

FIG. 14 shows a flow diagram for a burst interleaving technique for fullduplex impulse radio communications, in accordance with an embodiment ofthe present invention.

FIG. 15 shows exemplary pulses for a further embodiment of the presentinvention using different pulse repetition frequencies for twocommunicating transceivers.

FIG. 16 illustrates the cross correlation process in accordance with thepresent invention.

FIG. 17 shows a representative illustration of an impulse radiotransceiver for full duplex communications, in accordance with anembodiment of the present invention.

FIG. 18 shows a representative illustration of an impulse radiotransceiver for full duplex communications, in accordance with anotherembodiment of the present invention.

FIG. 19 shows an exemplary block diagram of a transceiver implementedfor synchronizing pulse interleaving, according to a preferredembodiment of the present invention.

FIG. 20 shows a flow diagram to implement a delay for pulse interleavedcommunications.

FIG. 21 illustrates acquisition using a conventional method of a slidingcorrelation.

FIG. 22 shows misalignment of two time bases in accordance with thepresent invention.

FIG. 23 shows a representative block diagram of an impulse radioreceiver for fast lock, in accordance with the present invention.

FIG. 24 shows an exemplary block diagram for the extra-count logic ofthe receiver in FIG. 23.

FIG. 25 illustrates pulse width tau (τ) and frame (F) length of amonocycle pulse.

FIG. 26 shows a flow diagram illustrating operation of signalacquisition and lock according to the present invention.

In the drawings, like reference numbers indicate identical orfunctionally similar elements. Additionally, the left-most digit of thereference number identifies the drawing in which the reference numberfirst appears.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Table of Contents

I. Overview . . . 11

II. Technology Basics . . . 12

-   -   A. Gaussian Monocycle . . . 13    -   B. A Pulse Train . . . 15    -   C. Modulation . . . 15    -   D. Coding for Energy Smoothing and Channelization . . . 16    -   E. Reception and Demodulation . . . 17    -   F. Jam Resistance . . . 18    -   G. Processing Gain . . . 18    -   H. Capacity . . . 19    -   I. Multipath and Propagation . . . 20

II. Full Duplex for Impulse Radio Communication Systems . . . 22

-   -   A. The Impact of the Width of the Dither Window on System        Performance . . . 28

IV. Exemplary Transceiver Hardware . . . 29

-   -   A. Transmitter . . . 29    -   B. Receiver . . . 30    -   C. Time Hand-off . . . 32    -   D. Differential Rate Duplex . . . 33

V. Other Considerations . . . 34

VI. Fast Locking Mechanism for Channelized Ultrawide-band Communications. . . 35

VII. Fast Locking Analysis and Operation . . . 39

VIII. Conclusion . . . 42

I. Overview

A new technology-called ultra-wideband radio-promises to helpovercrowding of the radio spectrum. Unlike conventional wirelesssystems, which use narrowband modulated carrier waves to transmitinformation, ultra-wideband transmits over a wide swath of radiospectrum. Ultra-wideband may cause significantly less interference thanconventional narrowband radio solutions while safely coexisting withother wireless technologies on the market. Thus, ultra-wideband systemsmay allow a number of devices to share the same radio spectrum, helpingto alleviate the crowding of the radio spectrum. One embodiment ofultra-wideband radio is an impulse radio as described in more detailbelow. While embodiments of the present invention have been described inconjunction with impulse radio, it will, however, be generallyunderstood that embodiments of the present invention will apply equallywell to virtually all other types of ultra-wideband radios and systems.

Impulse radios generally have: short duration pulses; center frequenciestypically between 50 MHz and 10 gigahertz (GHz); ultrawide bandwidths of100+% of the center frequency; multi-mile ranges with sub-milliwattaverage power levels, even with low gain antennas; extremely low powerspectral densities; lower cost than other sophisticated radio designs,especially spread spectrum systems; and excellent immunity to jammingfrom other systems and to multipath fading.

Impulse radios have exceptional multipath immunity and are relativelysimple and less costly to build, especially in comparison to spreadspectrum radios. Impulse radio systems consume substantially less powerthan existing conventional radios. Additionally, impulse radio systemsoccupy less space than existing portable telecommunicationstransceivers. Because of these characteristics, impulse radio is anoptimal technology for a wide variety of applications, includingpersonal communications systems and in-building communications systems.

Copending, commonly assigned U.S. patent application Ser. No.08/309,973; U.S. Pat. No. 5,677,927 (filed Sep. 20, 1994, and titled AnUltrawide-Band Communication System and Method; which is incorporatedherein by reference and referred to as the '973 application) describesthe following impulse radio features: the use of impulse radiosubcarriers; the time modulator that is used for code time delaying andsubcarrier time delaying; linearization of the time modulator; pseudoManchester coding for modulation of digital data using impulse radiocommunications; and a lock acquisition scheme for the impulse radioreceiver to acquire and maintain lock of impulse radio signals. A fullduplex impulse radio system is described in copending, commonly assignedU.S. patent application Ser. No. 08/428,489; U.S. Pat. No. 5,687,163(filed Apr. 27, 1995, and titled Full Duplex Ultrawide-BandCommunication System and Method, which is also incorporated herein byreference).

Section II is directed to technology basics and provides the reader withan introduction to impulse radio concepts, as well as other relevantaspects of communications theory.

Section III is directed full duplex for impulse radio communicationsystems. This section includes subsections relating to the theory ofoperation of full duplex for an impulse radio transceiver.

Section VI is directed to embodiments of a fast locking mechanism forchannelized ultrawide-band communications.

II. Technology Basics

As stated above, this section is directed to technology basics andprovides the reader with an introduction to impulse radio concepts, aswell as other relevant aspects of communications theory. This sectionincludes subsections relating to Gaussian monocycle pulses, pulse trainsof gaussian monocycle pulses, modulation, coding, and qualitative andquantitative characteristics of these concepts.

Impulse radio transmitters emit short Gaussian monocycle pulses with atightly controlled average pulse-to-pulse interval. Impulse radiotransmitters use pulse widths of between 20 and 0.1 nanoseconds (ns) andpulse-to-pulse intervals of between 2 and 5000 ns. These narrowmonocycle pulses have inherently wide-band frequency characteristics.

Impulse radio systems uses pulse position modulation, with the actualpulse-to-pulse interval being varied on a pulse-by-pulse basis by twocomponents: an information component and a pseudo-random code component.Unlike spread spectrum systems, the pseudo-random code is not necessaryfor energy spreading (because the impulses themselves are inherentlywide-band), but rather for channelization, energy smoothing in thefrequency domain, and jamming resistance.

The impulse radio receiver is a direct conversion receiver with a crosscorrelator front end. The front end coherently converts theelectromagnetic pulse train to a baseband signal in one stage. Theimpulse radio receiver integrates multiple pulses to recover each bit ofthe transmitted information.

A. Gaussian Monocycle

The most basic element of impulse radio technology is the practicalimplementation of Gaussian monocycles, which are also referred to hereinas Gaussian monocycle pulses. A Gaussian monocycle is the firstderivative of the Gaussian function. FIGS. 1A and 1B show a 2 GHz centerfrequency (i.e., a 0.5 ns pulse width) monocycle pulse in the time andfrequency domains (see 102 and 104, respectively). (Actual practiceprevents the transmission of a perfect Gaussian monocycle. In thefrequency domain, this results in a slight reduction in the signal'sbandwidth.) These monocycles, which are sometimes called impulses, arenot gated sine waves.

The Gaussian monocycle waveform is naturally a wide bandwidth signal,with the center frequency and the bandwidth completely dependent uponthe pulse's width. In the time domain, the Gaussian monocycle isdescribed mathematically by:${V(t)} = {A\frac{\sqrt{2e}}{\tau}t\quad{\mathbb{e}}^{- {(\frac{t}{\tau})}^{2}}}$

Where, A is the peak amplitude of the pulse, t is time, and τ (tau) is atime decay constant.In the frequency domain, the Gaussian monocycle envelope is:${V(\omega)} = {A\quad{\omega\tau}^{2}\sqrt{2\pi\quad e}{\mathbb{e}}^{\frac{\omega^{2}\tau^{2}}{2}}}$The center frequency is then: ${fc} = {\frac{1}{2{\pi\tau}}\quad{Hz}}$Relative to c, the 3 dB down points (power) are:f_(lower)=0.319c; f_(upper)=1.922c.

Thus, the bandwidth is approximately 160% of the center frequency.Because τ (tau) also defines the pulse width, then the pulse widthspecifies both the center frequency and bandwidth. In practice, thecenter frequency of a monocycle pulse is approximately the reciprocal ofits length, and its bandwidth is approximately equal to 1.6 times thecenter frequency. Thus, for the “0.5 ns” pulse shown in FIGS. 1A and 1B:f_(c)=2.0 GHz; Δf_(c)=3.2 GHz.

B. A Pulse Train

Impulse radio systems use pulse trains, not single pulses, forcommunications. As described in detail in Section III below, the impulseradio transmitter produces and outputs a train of pulses for each bit ofinformation.

Prototypes built by the inventors have pulse repetition frequencies ofbetween 0.7 and 10 megapulses per second (mpps, where each megapulse is10⁶ pulses). FIGS. 2A and 2B are illustrations of a 1 mpps system with(uncoded, unmodulated) 1 ns pulses in the time and frequency domains(see 102 and 104, respectively). In the frequency domain, this highlyregular pulse train produces energy spikes (comb lines 204) at onemegahertz intervals; thus, the already low power is spread among thecomb lines 204. This pulse train carries no information and, because ofthe regularity of the energy spikes, might interfere with conventionalradio systems at short ranges.

Impulse radio systems have very low duty cycles so the average power inthe time domain is significantly lower than the peak power in the timedomain. In the example in FIGS. 2A and 2B, for example, the impulsetransmitter operates 0.1% of the time (i.e., 1 ns per microsecond (ps)).

Additional processing is needed to modulate the pulse train so that theimpulse radio system can actually communicate information. Theadditional processing also smoothes the energy distribution in thefrequency domain so that impulse radio transmissions (e.g., signals)interfere minimally with conventional radio systems.

C. Modulation

Amplitude and frequency/phase modulation are unsuitable for thisparticular form of impulse communications; the only suitable choice ispulse position modulation, which allows the use of a matched filter(i.e., cross correlator) in the receiver. As illustrated in FIG. 3, amodulating signal changes the pulse repetition interval (PRI) inproportion to the modulation.

If the modulating signal were to have three levels, the first levelmight shift the generation of the pulse forward in time from the nominalby □ picoseconds (ps); the second level might not shift the pulseposition in time from the nominal at all; and the third level mightdelay the pulse by □ ps. This would be a digital modulation scheme.Analog modulation would allow continuous deviations between PRI−□ andPRI+□. In the impulse radio system the maximum value of □ is t/4, wheret=time of the pulse. The time measurement is assumed to be taken fromthe same part of the monocycle waveform on successive monocycles.

In the frequency domain, pulse position modulation distributes theenergy over more frequencies. For example, in the case of a 1 mppssystem where the modulation dither (d) is 100 ps, the PRI is 1,000,000Hertz (Hz) and the additional frequency components are: 999,800.04 Hz,999,900.01 Hz, 1,000,100.01 Hz, and 1,000,200.04 Hz. (Dither is animpulse radio communications term for moving the position of a pulse intime.) Transmitted energy is now distributed among more spikes (comblines) in the frequency domain. If the total transmitted energy remainsconstant, the energy in each frequency spike decreases as the number ofpossible pulse positions increases. Thus, in the frequency domain, theenergy is more smoothly distributed.

D. Coding for Energy Smoothing and Channelization

Because the receiver is a cross correlator, the amount of time positionmodulation required for one-hundred percent modulation is calculated bythe inverse of f_(c)/4 (where f_(c) is the center frequency). For amonocycle with a center frequency of 1.3 GHz, for example, thiscorresponds to ±57 (ps) of time position modulation. Thespectrum-smoothing effects at this level of time dither is negligible.

Impulse radio achieves optimal smoothing by applying to each pulse a PNcode dither with a much larger magnitude than the modulation dither.FIG. 4 is a plot illustrating the impact of pseudo-random dither onenergy distribution in the frequency domain. FIG. 4, when compared toFIG. 2B, shows the impact of using a 256 position PN code relative to anuncoded signal.

PN dithering also provides for channelization (channelization is aprocedure employed to divide a communications path into a number ofchannels). In an uncoded system, differentiating between separatetransmitters would be very hard. PN codes create channels, if the codesthemselves are relatively orthogonal (i.e., there is low correlationand/or interference between the codes being used).

E. Reception and Demodulation

Clearly, if there were a large number of impulse radio users within aconfined area, there might be mutual interference. Further, while theuse of the PN coding minimizes that interference, as the number of usersrises the probability of an individual pulse from one user's sequencebeing received simultaneously with a pulse from another user's sequenceincreases. Fortunately, implementations of an impulse radio according tothe present invention do not depend on receiving every pulse. Theimpulse radio receiver performs a correlating, synchronous receivingfunction (at the RF level) that uses a statistical sampling of manypulses to recover the transmitted information.

Impulse radio receivers typically integrate 200 or more pulses to yieldthe demodulated output. The optimal number of pulses over which thereceiver integrates is dependent on a number of variables, includingpulse rate, bit rate, jamming levels, and range.

F. Jam Resistance

Besides channelization and energy smoothing, the PN coding also makesimpulse radio highly resistant to jamming from all radio communicationssystems, including other impulse radio transmitters. This is critical asany other signals within the band occupied by an impulse signal act as ajammer to the impulse radio. Since there are no unallocated 1+GHz bandsavailable for impulse systems, they must share spectrum with otherconventional and impulse radios without being adversely affected. The PNcode helps impulse systems discriminate between the intended impulsetransmission and transmissions from others.

FIG. 5 illustrates the result of a narrowband sinusoidal jamming(interference) signal 502 overlaying an impulse radio signal 504. At theimpulse radio receiver, the input to the cross correlator would includethat narrowband signal 502, as well as the received ultrawide-bandimpulse radio signal 504. Without PN coding, the cross correlator wouldsample the jamming signal 502 with such regularity that the jammingsignals could cause significant interference to the impulse radioreceiver. However, when the transmitted impulse signal is encoded withthe PN code dither (and the impulse radio receiver is synchronized withthat identical PN code dither) it samples the jamming signals randomly.According to the present invention, integrating over many pulses negatesthe impact of jamming.

In statistical terms, the pseudo-randomization in time of the receiveprocess creates a stream of randomly distributed values with a mean ofzero (for jamming signals). Therefore, to eliminate the impact ofjammers all that is necessary is to sample over enough pulses (i.e.,integrate over a sufficiently large number of pulses) to drive theimpact of the jamming signals to zero.

G. Processing Gain

Impulse radio is jam resistant because of its large processing gain. Forspread spectrum systems, the definition of processing gain, whichquantifies the decrease in channel interference when wide-bandcommunications are used, is the ratio of the bandwidth of the channel tothe bandwidth of the information signal. For example, a direct sequencespread spectrum system with a 10 kHz information bandwidth and a 16 MHzchannel bandwidth yields a processing gain of 1600 or 32 dB. However,far greater processing gains are achieved with impulse radio systemswhere, for the same 10 kHz information bandwidth and a 2 GHz channelbandwidth, the processing gain is 200,000 or 53 dB.

The duty cycle (e.g., of 0.5%) yields a process gain of 28.3 dB. (Theprocess gain is generally the ratio of the bandwidth of a receivedsignal to the bandwidth of the received information signal.) Theeffective oversampling from integrating over multiple pulses to recoverthe information (e.g., integrating over 200 pulses) yields a processgain of 28.3 dB. Thus, a 2 GHz divided by a 10 mpps link transmitting 50kilobits per second (kbps) would have a process gain of 49 dB, (i.e.,0.5 ns pulse width divided by a 100 ns pulse repetition interval wouldhave a 0.5% duty cycle, and 10 mpps divided by a 50,000 bps would have200 pulses per bit.)

H. Capacity

Theoretical analyses suggests that impulse radio systems can havethousands of voice channels per cell. To understand the capacity of animpulse radio system one must carefully examine the performance of thecross correlator. FIG. 6 shows the “cross correlator transfer function”602. This represents the output value of an impulse radio receiver crosscorrelator for any given received pulse. As illustrated at 604, thecross correlator's output is 0 volts when pulses arrive outside of across correlation window 606. As a received pulse 608 slides through thewindow, the cross correlator output varies. It is at its maximum (e.g.,1 volt) when the pulse is T/4 ahead of the center of the window (asshown at 610), 0 volts when centered in the window (as shown at 612);and at its minimum (e.g., −1 volt) when it is T/4 after the center (notshown).

When the receiving system is synchronized with the intended transmitter,the cross correlator's output has a swing of between ±1 volt (as afunction of the transmitter's modulation). Other in-band transmissionwould cause a variance to the cross correlator's output value. Thisvariance is a random variable and can be modelled as a Gaussian whitenoise signal with a mean value of 0. As the number of interferersincreases, the variance increases linearly. By integrating over a largenumber of pulses, the receiver develops an estimate of the transmittedsignal's modulation value. Mathematically:${{Variance}\quad{of}\quad{the}\quad{Estimate}} = \frac{N\quad\sigma}{\sqrt{Z}}$Where N=number of interferers, σ is the variance of all the interferersto a single cross correlation, and Z is the number of pulses over whichthe receiver integrates to recover the modulation.

This is a good relationship for a communications system, for as thenumber of simultaneous users increases, the link quality degradesgradually (rather than suddenly).

I. Multipath and Propagation

Multipath fading, the bane of sinusoidal systems, is much less of aproblem (i.e., orders of magnitude less) for impulse systems than forconventional radio systems. In fact, Rayleigh fading, so noticeable incellular communications, is a continuous wave phenomenon, not an impulsecommunications phenomenon.

In an impulse radio system, in order for there to be multipath effectsspecial conditions must persist. First, the path length traveled by thescattered pulse must be less than the pulse's width times the speed oflight. Second, successively emitted pulses at the transmitter may arriveat the receiver at the same time neglecting the decorrelation benefitsof time coding.

For the former (with a one nanosecond pulse), that equals 0.3 meters orabout 1 foot (i.e., 1 ns×300,000,000 meters/second). (See FIG. 7, in thecase where the pulse traveling “Path 1” arrives one half a pulse widthafter the direct path pulse.)

For the latter (with a 1 megapulse per second system), that would beequal to traveling an extra 300, 600, 900, etc. meters. However, becauseeach individual pulse is subject to the pseudo-random dither, thesepulses are decorrelated.

Pulses traveling between these intervals do not cause self-interference(in FIG. 7, this is illustrated by the pulse traveling Path 2). However,pulses traveling grazing paths, as illustrated in FIG. 7 by thenarrowest ellipsoid, create impulse radio multipath effects.

As illustrated in FIG. 8 at 802, if the multipath pulse travels one halfwidth of a pulse width further, it increases the power level of thereceived signal (the phase of the multipath pulse will be inverted bythe reflecting surface). If the pulse travels less than one half a pulsewidth further, it will create destructive interference as shown at 804.For a 1 ns pulse, for example, destructive interference will occur ifthe multipath pulse travels between 0 and 15 cm (0 and 6 inches).

Tests of impulse radio systems (including impulse radar tests) suggestthat multipath will not present any major problems in actual operation.Additionally, shorter pulse widths are also envisioned, which willfurther reduce the probability of destructive interference (because thereflected path length required for destructive interference will beshortened).

III. Full Duplex for Impulse Radio Communication Systems

A representative block diagram of a full duplex impulse radiocommunication system is shown in FIG. 9. A first transceiver (A) 902comprises a transmitter (T1) 904 and a receiver (R1) 906. A secondtransceiver (B) 908 comprises a transmitter (T2) 910 and a receiver (R2)912. The transceivers 902 and 908 are separated by a propagation medium914, such as air, space, or other medium cable of propagatingultrawide-band signals. Transmitted impulse radio signals 916 propagatethrough the propagation medium 914 between T1 904 and R2 912, andbetween T2 910 and R1 906.

The purpose for full duplex transmission in an ultrawide band impulseradio system is to provide two-way transmittal of information similar totelephony, as opposed to a walkie-talkie (i.e., a push-to-talk simplexoperation). Since ultrawide band signals utilize the fullelectromagnetic spectrum, or at least a very large part of it, it isnecessary to use some technique other than frequency domain duplexing,which is the conventional method. The inventors have therefore developeda pulse interleaving technique for full duplex impulse radiocommunications.

For example, with reference to FIG. 10, if transmitter T1 904 sends outa train of modulated pulses 1002, receiver R1 906 would need to receivepulses 1004 transmitted from transmitter T2 910 during the time periodsbetween the pulses 1002 transmitted by T1.

One complication with this implementation is that at certain integralranges between transmitter/receiver pair number 1 (i.e., transceiver 1and transceiver 2), it will be necessary for one or the other totransmit and receive exactly simultaneously. However, simultaneoustransmission and reception requires too large of a dynamic range in thereceiver to allow functionality. This means that at certain discretelocations, determined by the pulse repetition rate, it will be necessaryfor each transceiver to transmit and receive simultaneously. As shown inFIG. 11, pulses 1102 transmitted by T1 904 and pulses 1104 transmittedby T2 910 pass exactly on top of each other at positions calledcontention zones. There will be a series of these contention zones,which cannot be practically removed. Even if one or both transceiversare mobile, as they move with respect to each other, they will stillcreate contention zones.

According to an embodiment of the present invention, T1 904 is set toemit each pulse 1202 10 nanoseconds (ns) after R1 906 receives a pulse1204 from T2 910. This transmit delay is depicted in FIG. 12. Thisreduces interference between the transmitter and the receiver attransceiver 1, for example. If T1 904 transmits after it receives apulse, those pulses cannot interfere. Since T1 904 has waited for over awhole period (one period is about 5 ns) before transmitting, most of thenoise from the previous pulse has died down before the current pulse istransmitted. However, some contention zones 1206 will still existbetween the two transmitters.

The easiest way to resolve these contention zones 1206 is to permit thefirst transceiver to have a choice of say, 10 ns or 100 ns of delaybefore transmitting after receiving a pulse. This removes theinterference at point 1208 for example, by pushing (position in time)pulse 1210 up to point 1212 so that the self-interference is avoided.

In addition, it is important to remember that in all cases, each pulseis also time dither coded as described above. They are shown here asun-time dither coded for simplicity. Thus, time dither coding furtherserves to remove the interface 1208.

The steps required in signal acquisition for pulse interleaving areshown in a flow diagram in FIG. 13. In operation, T1 904 would begintransmitting to R2 912, as shown at a step 1302. R2 912 scans fordetection and acquires lock through its scanning mechanism (see step1304). Once it 1304). Once it acquires lock (see step 1306), itsaccompanying transmitter (T2 910) can begin transmitting, as shown at astep 1308. R1 906 then scans for detection, at step 1310. If R1 906happens to be in a contention zone, then it will never acquire lock toT2 910. Therefore, at the message level, R1 906 must wait for anacknowledge message (ACK) 1306 to be conveyed to it by T1 904 before itknows whether to use the 10 ns or the 100 ns transmitter receive timingdelay. If it never receives, or after a certain time does not receivethe ACK that R1 906 has acquired T2 910, then T2 910 times-out andshifts its transmitted pulse timing by 100 ns, for example, and triesagain. These steps are shown generally by a conditional loop at steps1312, 1314, 1316 and 1318.

Whereupon if R2 912 does acqliire lock (i.e., receives an ACK from T1904 sent at step 1320) as shown at step 1322, T2 910 will transmit areturn ACK at step 1324, a link is established, and the transceivers arein lock.

The timeout is preferably the maximum time period required for R2 912 toscan for a pulse from T1 904 over the entire modulo of the dither code.For a 256 bit code, and a fairly small code dither of 10 ns a timeoutcan take up to 20 seconds. Timeout is only done for an initial lock. Atimeout is not needed if the transceivers switch codes or delay values.Because of the simplicity in implementation of the pulse interleavetechnique, pulse interleave full duplex is very economical for manycommunication applications, such as telemetry and transponder-typesystems. In the preferred embodiment, the receiver can stay on so that acold start is not necessary.

As discussed above, the mobile environment presents unique contentionzone problems. Therefore, the following embodiments deal with the mobileenvironment explicitly, and are specifically directed at providingimmunity to dead or contention zone problems.

One embodiment of the present invention directed to these problems is aburst interleave method. According to the burst interleave method, thereis no contention at all. The burst interleave method is shown in a flowdiagram in FIG. 14. T1 904 starts the process by transmitting a burst(see step 1402), which, for example, could be 10 microseconds in length.

In an exemplary embodiment, each burst contains 20 pulses at a 2megapulse per second rate, or 50 pulses at a 5 megapulse rate. Thisfirst transmitted burst is received by R2 912 after a certain amount oftime passes due to propagation delay (i.e., range delay) and scanningdelay by R2 912 (see step 1404). Range delay corresponds to about 5.2microseconds per mile (approximately 5,200 feet) or about one foot pernanosecond.

At the end of this received burst, R2 acquires lock (see step 1406) andthen T2 910 transmits its burst containing information modulation (atstep 1408), and after the same range delay, R1 scans for detection (step1410) and acquires lock (step 1412). If the timing between the bursts issufficient, then under no circumstance(s) of position or range betweenthe transceivers do the bursts collide. The criterion is that the delaybetween bursts be sufficient to accommodate the round trip delay andburst width. In practice, the burst should be as far away as possiblebefore using up all the margin of receive time in this receiver beforeit will be required to transmit again. The transceivers then swapacquisition messages, as shown at steps 1414, 1416, 1418 and 1420, tocomplete the locking process.

A further embodiment of the present invention uses code divisionmultiple access (CDMA) for achieving full duplex communication in anultrawide band impulse radio system. In this variation T1 904 and T2 910are operated with different time dither codes, with dither windowsnearly equalling the full frame so that each successive pulse can appearanywhere within the period separating the pulses. (The dither window isthe period within which a monocycle can occur when position modulated bya dither code.) T1 904 and T2 910 can even use the same dither codebecause a time delay between them permits decorrelation. Typically,however, they would be operated on different time dither codes.

In this embodiment, T1 904 generates a blanking pulse that preventsreceiving any energy within a certain amount of time after transmission,for example, 10 ns. This allows the antennas in the local environment toring down or dampen energy for opening the receiver for possiblereceived pulse. For example, a pulse width of 0.5 ns (or centerfrequency of 2 gigahertz), with a period of 200 ns (which is therepetition rate of 5 megapulses per second), produces a cycle of 1 in400 (i.e., 0.25%).

A blanking pulse equalling the transmitted pulse emitted is, however,not entirely effective. There is still sufficient energy ringing down inthe environment and in the antenna that may cause significantself-interference. Statistically, pulses can align themselves perfectlyin only about 1 in 400 pulses. The blanking window of 10 ns increasesthe probability of a received pulse being within that blanking window,up to 1%. A 1% probability means that 1% of the energy is thrown away bythe receiver. A loss of only 1% of transmitted energy is a very smallpenalty to exact to allow for a full duplex operation. This 1% reductionlikely unmeasurable.

A still further embodiment is frequency division multiple access (FDMA),where the word “frequency” stands for pulse repetition frequency, whichdistinguishes this term from that used in continuous wave FM systems.

FIG. 15 shows exemplary pulses for this embodiment, in which T1 904 isoperated for example, at 1 megapulse per second (represented bymicrosecond pulses 1502 (numbers 1, 2, 3, 4, 5, 6 and so on). AssumingT2 910 is operating on, about 0.85 microseconds per period (see pulses1504), after six pulses the two will come into alignment and beapproximately settled. But after that time, however, all of the pulsesmiss.

Therefore, if the timed coding is confined to a relatively narrow window(say 4 ns, which is used for a 2 gigahertz center frequency system) thenno matter what the placement of the two transceivers relative to eachother, only one in six pulses will collide with each other. In practice,the repetition rate difference between the two would be such that onlyone in a hundred would cause a collision 1506. That one in a hundred canbe blanked out (similar to the preceding example), which would againcause a 1% in reduction in power available to the receiver.

Blanking can be implemented in many ways. Discrete logic can be used todetermine when received pulses and transmitted pulses of two differentpulse repetition rates will interfere or occur too close in time.Interference is avoided by gating off one of the trigger signals (forexample).

This FDMA embodiment has some of the advantages of the pulseinterleaving embodiment, such as 100% availability of the transmitter.The pulse interleaving embodiment requires the transmitter to be turnedoff for a significant fraction at the transmitting cycle. Thedisadvantage being, for the same average of transmitted power, the pulsepower has to be that much higher to make up for it. The duty cycle inthe first example was on the order of 33%. Therefore the pulse power(i.e., the instantaneous pulse power), would have to be 66% larger. Thislast embodiment shares the advantages of pulse interleave—100%availability of the carrier—but it is never turned off on transmit. Onreceiving however, the periodic self-interference is taken care of byblanking, as in the previous example, reducing the received poweravailability by only 1%, a perfectly acceptable number.

The method used to provide for isolation between a transmitter and areceiver for a full duplex impulse radio link is different than forconventional radios because conventional radios operate using continuouswave carrier frequencies. These carrier frequencies can be verynarrow-band and as such, frequency domain techniques can be used toisolate the transmitter from the receiver in the same view. Low passfilters can be used on the transmitter to prevent spurious energy fromgetting into a receiver, which is operated at a slightly higherfrequency. Conversely, a high pass filter is used to eliminate powerfrom the transmitter from getting into the receiver. This conventionalfiltering, however, cannot effectively be applied to impulse radiosystems because the transmitter and receiver use the same pulse withmonocycle.

The operating characteristics of an impulse radio system thereforerequire a different isolation/filtering approach. This can best beillustrated by way of example. Two monocycles pulses with differentcenter frequencies are shown in FIG. 16. A long monocycle 1602 has a lowfrequency content, and a shorter monocycle 1604 has a higher centerfrequency. Although these two pulses differ in center frequency bynearly 3 to 1, they still significantly overlap. Therefore, even in thiscase a filter can be used to provide some isolation between atransmitter and a receiver, operating at one center frequency (f_(c1))on the uplink and different center frequency (f_(c2)) on the downlink.In this embodiment contention is completely eliminated by the fact thatdifferent center frequencies are used in operation.

A. The Impact of the Width of the Dither Window on System Performance

As note above, the dither window is the period within which a monocyclecan occur as positioned by a dither code. In the above examples, thedither window is 5 ns wide. Each dither window is separated by 200 ns.Thus, a subsequent monocycle can occur anywhere within the next ditherwindow, and at a minimum, 200 ns later. The concentration of pulses in arelatively narrow time zone in each frame, where a frame is the nominalinterpulse interval, contributes to increased interference withconventional services, as well as increased interference with liketransceivers. The increased interference is an undesirable consequenceof the difficulty of making wider dither windows.

The difficulty lies in the fact that long time delays are difficult tomake with low jitter. Because this is a coherent communication scheme,low jitter is important for efficient conversion of a pulse and for goodsignal-to-noise ratio at low RF power levels.

The pulse interleave method, burst interleave method, and the pulserepetition rate multiple access techniques are all three consequences ofthis concentration of energy in a small time zone. As this window iswidened, the constraints are less on the system until at a limit, awhole frame can be a target for a gain given monocycle (i.e., in a 200ns average pulse rate, a pulse can appear anywhere within that 200 ns).For generality sake, a brief off-time between dither windows isdesirable.

In the pulse interleave, burst interleave, CDMA and the repetition ratemultiple access techniques, the distinction between all these types ofinterleaves disappears at the full frame. They are indistinguishablefrom one another. This is because once the structure as removed by fullframe dither, further shuffling cannot make it any more random. Inaddition, interleaving will not work when there are no quiet gaps.

IV. Exemplary Transceiver Hardware

A. Transmitter

A preferred embodiment of an impulse radio transmitter 904 or 910 of animpulse radio communication system will now be described with referenceto FIG. 17.

The transmitter 1700 comprises a time base 1702 that generates aperiodic timing signal 1704, which is provided to a time delay modulator1706. The time delay modulator 1706 modulates the periodic timing signal1704 with an information signal 1708 from an information source, to togenerate a modulated timing signal 1710. The modulated timing signal1710 is provided to a code time modulator 1712 that dithers themodulated timing signal 1710 using a pseudo noise code. The code timemodulator 1712 outputs a modulated, coded timing signal 1714 to anoutput-stage 1716. The output stage 1716 uses the modulated, codedtiming signal 1714 as a trigger to generate electrical monocycle pulses(not shown). The electrical monocycle pulses are sent to a transmitantenna 1718 via a transmission line 1720 coupled thereto. Theelectrical monocycle pulses are converted into propagatingelectromagnetic pulses 1722 by the transmit antenna 1718. A detaileddescription of various impulse radio transmitters is included in the'973 application.

B. Receiver

An impulse radio receiver 1701 will now described with reference to FIG.17. An impulse radio receiver (hereafter called the receiver) 1701comprises a receive antenna 1726 for receiving a propagated impulseradio signal 1724. A received signal is input to a cross correlator 1728via a receiver transmission line 1730, coupled to the receive antenna1726.

The receiver 1701 also comprises a decode timing modulator/decode source1732 and an adjustable time base 1734. (The adjustable time base 1734can comprise a voltage controlled oscillator or a variable delaygenerator, as would be apparent to a person skilled in the art.) Thedecode timing modulator/decode source 1732 (hereafter called the decodetiming modulator) generates a decode signal 1736 corresponding to the PNcode used by the associated impulse radio transmitter (not shown) thattransmitted the propagated signal 1724. The adjustable time base 1734generates a periodic timing signal 1738 that comprises a train oftemplate signal pulses having waveforms substantially equivalent to eachpulse of the received signal 1724.

The detection process performed by the cross correlator 1728 comprises across correlation operation of the received signal 1724 with the decodesignal 1736. Integration over time of the cross correlation generates abaseband signal 1740. The baseband signal 1740 is demodulated by ademodulator 1742 to yield a demodulated information (signal) 1744. Thedemodulated information signal 1744 is substantially identical to theinformation signal of the transmitter that sent the received signal1724.

The baseband signal 1740 is also input to a lowpass filter 1746. Thelowpass filter 1746 generates an error signal 1748 for an acquisitionand lock controller 1750 to provide minor phase adjustments to theadjustable time base 1734. A detailed description of an impulse radioreceiver is included in the '973 application.

FIG. 18 is a transceiver block diagram for the burst interleaveembodiment of the present invention. A transmitter burst controller 1802and a receiver burst controller 1804 are added to the basic architectureof the transceiver of FIG. 17. These two controllers are state machinesthat can be hardwired or programmably controlled (using EEPROMS, or thelike) to time position the modulated, coded timing signal 1714 and totime modulate the periodic timing signal 1738, respectively, inaccordance with the burst interleave operation described above.

The delay required for the pulse interleave embodiment of the presentinvention is determined and provided by the acquisition and lockcontroller 1750. Similarly, for the other embodiments, the pulserepetition rate, dither window and are hardwired or programmablycontrolled into the burst controllers 1802, 1804 and the acquisition andlock controller 1750, for example. Other control features andmodifications to the disclosed transceiver components/controllers wouldbe apparent to a person skilled in the relevant art without departingfrom the scope of the present invention.

C. Time Hand-off

For the pulse interleave embodiment, each receiver must measure the timebetween the reception of a pulse from another transceiver and thetrigger to its own transmitter (this which can be accomplished withconventional circuitry). When one transceiver detects that this time isbelow a minimum limit (e.g., 20 ns), it notifies the other transceiverto synchronously change its receive timing (and the first transceiverwill change its transmit timing) at, for example, the first pulse of thesecond code modulo from now. Where “now” is a point in time determinedby the first transceiver as a reference point in time that iscommunicated to the second transceiver (or otherwise inferred by thesecond transceiver) for synchronization.

This is possible because, although it is not possible to “tag”individual pulses using modulation (since many pulses make up a bit),modulos are long enough to encode at least one whole bit, and thattherefore can serve as a trigger for the counting of whole modulos.Since the coder keeps track of the pulse “count” in order to apply thecorrect time dither to the decoder, this method can indirectly identifyindividual pulses for the purpose of synchronization.

This process will be repeated any time the minimum time separation isdetected, which happens every 54.86 meters (180 feet) of travel at a 5MPPS rate for example.

A mechanism to accomplish the synchronization and locking for operationof pulse interleave can be discrete logic, but can be readilyimplemented by a digital signal processor (DSP) with minimal programmingthat would be apparent to a person skilled in the relevant art based onthis disclosure of the pulse interleave functionality.

FIG. 19 shows an exemplary block diagram of a transceiver implementedusing a DSP for synchronizing pulse interleaving, according to apreferred embodiment of the present invention. This figure shows enoughdetail of a transceiver to describe the synchronization. A DSP 1902 isused to determine whether the transmitter trigger signal 1904 is tooclose to the receiver trigger signal 1906, using a block 1908, labeled“measure time difference.” The DSP 1902 delays the transmitter triggersignal 1904 by 100 ns (for example) by sending a delay control signal1910 to a delay block 1912 to outputs a delayed trigger signal 1914,which is provided to the transmitter. The DSP 1902 also outputsmassaging information 1916 to be modulated with the data to accomplishthe synchronization with the other transceiver. A analog-to-digital(A/D) converter is shown at 1918, because the DSP need to process thecross correlator output in the digital domain.

FIG. 20 shows a flow diagram of the DSP operation to implement a delayfor pulse interleaved communications. From a cold start 2002, thetransceivers acquire lock 2004, as described above. If a time (t)between a transmitted pulse and a received pulse is less than 20 ns, asshown at a decisional block 2006, a 100 ns delay is negotiated betweenthe two transceivers at 2008. This is termed a negotiation, since eithertransceiver can perform the necessary delay. The negotiation is carriedout via massaging 1916. If lock is lost, as determined by decisionalblock 2010, acquisition must be repeated, as shown at 2012.

D. Differential Rate Duplex

In the pulse repetition rate embodiment, if the transmitter and receivercomprising a transceiver are operated at two different rates, then it isnot possible to “interleave” the pulses, since they “beat” with eachother (i.e., the timing of the pulse trains will periodically cause thetransmitted and received pulses to periodically coincide).

A mechanism similar to the detector described above can be used todetect the minimum pulse separation condition. However, this signal willbe employed in a different way: either to blank the trigger to thecorrelator correlator or to the transmitter. Either response will havethe desired result of preventing self interference, but they havedifferent tradeoffs in a communications system.

If the transmitter is blanked, it will reduce the transmitted power andinterfere with the carrier which would be received by anothertransceiver, due to the gaps in the carrier which result from theblanking action. However, it increases the received power to the firsttransceiver, since it will not have to throw away the pulses which occurwithin this minimum separation window as would be the case if thereceiver is blanked instead.

V. Other Considerations

The communications methods described here have been observed to beusable not only using radio (electromagnetic) impulsive waveforms, butalso may use acoustic signals. The principle differences in the latterapproach are: (1) frequency of operation and (2) signal transmission.

The frequency of operation is primarily between a few tens of Hertz(e.g., pulses of a duration of several tens of milliseconds), up to afew hundred Megahertz (e.g., pulses with durations of a fewnanoseconds).

Acoustic transducers are employed in the acoustic approach rather thanthe antennas, which are used for the radio approach. The signalcharacteristics of the transducers are similar to the signalcharacteristics required by the antennas used in the radio approach inthat they must be capable of transmitting and/or receiving waveformswith bandwidths of □ 100% of the center frequency (without more than afew percent dispersion, and with good conversion gain). Transducers maybe made from a material called Kynar Film supplied by PennwaltCorporation in Valley Forge, Pa. The geometry of a transducer made fromthis type as would become apparent to a person skilled in the relevantart.

IV. Fast Locking Mechanism for Channelized Ultrawide-band Communications

FIG. 21 illustrates acquisition using a conventional method of a slidingcorrelation. This figure shows a short sequence of eight pulses(chips)/modulo with a chip frame period of 1 microsecond (μs). Here thereceiver is shown out of synchronization with the received pulse train.As shown in this figure, the monocycle pulse may occur anywhere in thechip frame due to dithering. The time difference between the receivedwaveform and the cross correlator are shown in FIG. 21 to differ by onlyabout 2.2 ps. The time scales illustrated in this figure are greatlyexaggerated. At the time scales shown, the monocycle pulses aresub-nanosecond waveforms and would be invisible. Furthermore, inreality, the chip modulo would be 256 or some higher power of 2. Furtherstill, the chip modulo may be a non-repeating code, or the like. FIG. 22shows misalignment of two time bases illustrated as blocks using alarger time scale than that shown in FIG. 21. Each block of eight unitsindicates the period of a code modulo (8 μs) and the smaller blocks arethe chip frame time, within which a single, time coded monocycle pulsewill occur. According to the present invention, the interpulse period ofthe correlator is initially set to be slightly different than that ofthe received waveform, which is shown at the left hand side of thisfigure to be longer. The receiver's correlator comes into alignment atapproximately the 64 μs mark and thereafter maintains synchronizationusing feedback to adjust the correlator period to match that of thereceived waveform.

In the simple sliding lock technique of FIG. 21, the receiver'scorrelator PN time-hopping code progresses through its entire codemodulo at a rate slightly faster than the corresponding code generatorin a transmitter with which it is attempting to acquire a lock. Thisrate is determined by either the maximum offset frequency of anadjustable time base (typically a voltage control crystal oscillator orVCXO) or by the maximum rate of change of the frequency of theadjustable time base. Therefore, up to eight periods must be scannedpast each other in order to find the desired desired alignment. Forexample, an impulse radio transmitter operating at 1 mega (M) pulses persecond (pps) may be scanned by a receiver operating at a 20 ppm offset,which is a rate of 1/(20 Hz)=0.05 seconds per chip, where a chip isdefined as 1 monocycle pulse. In other words, a link using a code moduloof 250 pulses will take 12.5 seconds to be scanned. However, if thecenter frequency of the monocycle is 2 GHz, the bandwidth of thecorrelation signal that would be presented to the error circuit forlocking purposes will be 40 kHz. This is much too high a rate to controla typical VCXO, since such oscillators typically have a 1 kHz controlbandwidth.

The present invention, however, allows a receiver to lock to a(received) time dither coded signal in a minimum possible time, morequickly than can be accomplished by a simple sliding correlation searchas described in connection with FIG. 21. According to the presentinvention, the phase of the receiver adjustable time base isintentionally counted through its cycle with either an occasionalduplicated or dropped chip. This has the effect of jumping the phase ofthe receivers code generator one whole chip (for example) with respectto the transmitter's code generator without the necessity of sliding thecorrelator pulse to the next received pulse phase. While this is beingdone, the adjustable time base is also allowed to run either slightlyfaster or slower than the repetition rate of the transmitter, thusallowing the receiver's cross correlator to slide across the timebetween two pulses of the received signal. With the proper settings inthe receiver, all possible timing and code phases are examined duringthe drift from one pulse to the next.

The calculation of required dwell time of the receiver code phase isbased on the amount of energy contained in the received pulse.Generally, it is simply the same as the number of pulses used by thereceiver to assemble one bit, usually more than about 16 pulses, but inhigh noise environments this could require thousands of pulses. In thisway, the signal-to-noise ratio of the noise acquisition process will besimilar to that of the data recovery circuit in the receiver. Accordingto a preferred embodiment, an address counter driving a read only memory(ROM) containing the code table, is allowed to count successive timecode values for enough steps to allow that number of pulses to beintegrated to determine whether the current phase (code phase) is theproper one. Then the counter is either incremented or decremented by oneor more counts to slip the phase of the correlator. This process isrepeated continuously until all phases are tested at the current timeposition (pulse phase) or until coincidence of the received signal inthe template signal is detected. As noted above, an adjustable time basein the receiver is adjusted to allow the correlator to drift in pulsephase at the rate which allows all of the possible code phases to betested at each of the possible pulse phases.

A representative block diagram of an impulse radio receiver 2300 isshown in FIG. 23. Receiver 2300 receives impulse radio signals 2302propagated through a propagation medium (not shown) at an antenna 2304.A received signal 2306 is input to a cross correlator 2308 via areceiver transmission line 2310 coupled to the antenna 2302. A decodetiming modulator (dashed box) 2312 produces a decode signal 2314, whichis provided to the cross correlator 2308. The cross correlator 2308cross correlates the received signal 2306 with the decode signal 2314and outputs a baseband signal 2316. Once signal acquisition and lock aremade, as described below, the baseband signal 2316 is demodulated by ademodulator 2318, which outputs a demodulated information signal 2320.

The receiver 2300 also comprises an adjustable time base 2328. Theadjustable time base 2328 generates a periodic timing signal 2330. Acontroller 2332 generates a rate control signal 2334 to control the rateof the periodic timing signal 2330. The controller 2332 receives anerror signal 2336, which is a low pass filtered version of the basebandsignal 2316, via a low pass filter 2338.

Decode timing modulator 2312 comprises a (binary-to-time) delaygenerator 2322, a PN code and linearization read only memory (ROM) 2324,and an address counter and limit logic block 2326. Start address andstop address signals are provided to the address counter and limit logicblock 2326 from the controller 2332 via lines shown at 2340. Addressesare output from the address counter and limit logic block 2326 via a bus2327. The address counter and limit logic block 2326 provides addressesto access the PN code and linearization ROM 2324 when triggered by theperiodic timing signal 2330 provided by the adjustable time base 2328. APN code (that corresponds to a known PN code used by an impulse radiotransmitter) is output by the PN code and linearization ROM 2324 via abus 2325 and is provided to the (binary-to-time) delay generator 2322.The (binary-to-time) delay generator 2322 time modulates the periodictiming signal 2330 to generate the decode signal 2314.

Further details of delay generator 2322, read only memory (ROM) 2324 andaddress counter 2326 of the decode timing modulator 2312, as well as theoperation of the cross correlator 2308 and demodulator 2318 are fullydescribed in U.S. Pat. No. 5,677,927 and Ser. No. 08/428,489application, now U.S. Pat. No. 5,687,169, both of which are fullyincorporated by reference herein. For example, the adjustable time base2328 can comprise a programmable divider (not shown) and a voltagecontrolled oscillator (VCO) (not shown), which are used to output theperiodic timing signal 2330. A voltage control signal is provided to theVCO from the controller 2332 to adjust the VCO output, as will beapparent to a person skilled in the relevant art.

The cross correlator output is a wide band baseband signal (2316), whichis on the order of half the pulse repetition rate. For example, a 5 Mppsrate would yield a 2.5 MHz wide baseband signal (0-2.5 MHz). The sectionof that bandwidth that is of interest to the lock loop is in the kilohertz range and below. Therefore, the low pass filter 2338 cuts offfrequencies above about 10 kHz, unless a high speed lock process (i.e.,acquisition scheme) is employed, in which case 100 kHz may be thecutoff. Assuming that the controller 2332 is a microprocessor or adigital signal processor (DSP), such as a TMS320C40 DSP (manufactured byTexas Instruments, Dallas, Tex.), or the like, the high frequency doesnot affect the VCO (not shown) directly, and is easily handled by theDSP, which in turn controls the VCO.

Additional logic for acquisition and fast lock of impulse signalsincludes a counter 2342 that determines whether or not T chips have beenintegrated using the current code phase. If so, an extra count is addedusing discrete, extra-count logic 2344. Exemplary logic is shown in FIG.24. In this example, counter 2342 is a 16-chip counter that produces anoutput every T (16 for example) chips of the code modulo. The output2402 of the counter enables a one shot monostable timer 2404. The inputsof the counter 2342 and the monostable timer 2404 are triggered by theperiodic timing signal 2330. An output 2406 of the monostable timer 2404must be delayed by a delay element 2408 to avoid overlapping of itsoutput 2410 (called the “extra count”) with the periodic timing signal2330. The extra count output of the delay element is ANDed (via a gate2412) with the periodic timing signal 2330 and input to the addresscounter 2326.

Lock is detected via integration of T samples (see block 2350) andcomparison of a integration result 2352 via a threshold detector 2354.The threshold detector 2354 outputs a channel coincidence signal 2356 tothe controller 2332. Once coincidence is detected, the controller 2332disables the extra-count logic via a stop extra-count signal 2358, thusimplying signal lock.

VII. Fast Locking Analysis and Operation

FIG. 25 illustrates pulse width τ (tau) and frame length (F) (i.e., thepulse-to-pulse interval) of a monocycle pulse (not shown). Forexplanation and analysis of acquisition and fast locking according tothe present invention, the pulse width τ is subdivided into s samplingwindows (4 sampling windows are shown in the figure). A trial number (T)represents the number of pulses integrated by the integrator 2350 persampling window, prior to shifting the code count via the extra-countlogic 2344. The code's modulo length is M, which for this analysisM=256.

Given a center frequency of 2 GHz, the monocycle pulse width T is0.5×10⁻⁹ sec. for this example, and the frame width is 1×10⁻⁶ sec. Thetotal number of samples per frame is thus:$F_{s} = \frac{F \cdot s}{\tau}$

Using the above exemplary values, F_(s)=8000 samples. The worst casenumber of pulses to acquire code phase coincidence to result in a signallock is:F _(T) =F _(S) ·T·M

Which, using the above exemplary values, F_(T)=3.3×10⁷ pulses. Finally,the worst case time period to acquire code phase coincidence is:t ₁ =F _(T) ·F

Which, using the above exemplary values, t₁=32.8 sec. Modifications inoperation and/or hardware can be made to the lock mechanism to greatlyreduce this seemingly high value.

For example, the cross correlation rate can be sped-up by one or twoorders of magnitude. This will, however, increase the cost of the crosscorrelator. Alternatively, a plurality of less expensive crosscorrelators can be used in parallel. Each cross correlator in this casewould correlate a different section of the code, and the crosscorrelated results would need to be separately integrated for thresholddetection.

FIG. 26 shows a flow diagram illustrating operation of signalacquisition and lock according to the invention. From a cold start 2600,the receiver's controller 2332 starts sliding the rate of the adjustabletime base, as shown at 2602. One trial worth of correlation results areintegrated, at a step 2604, and received energy is compared to thethreshold, at a step 2606. If coincidence is detected (see “Yes” resultof conditional statement 2606) the controller stops sliding the timebase (at 2608) to maintain signal lock (at 2610). If the threshold isnot exceeded (see “No” result of conditional statement 2606), theaddress counter is incremented, as a step 2612.

The counter 2342 then determines whether T chips have sampled, at aconditional step 2614. If so, the extra-count logic in enabled and theaddress counter is incremented an extra code chip, as shown at step2616. If T chips have not yet been sampled, no extra count is added, andthe process returns to step 2604. A conditional step 2618 determineswhether all F_(T) pulses have been sampled. If so the acquisitionprocess is stopped, at step 2620, assuming there is no impulse signal todetect; otherwise, processing continues to step 2604.

In alternative embodiments, the step 2616 need not be a single chipincrement. The count can be incremented or decremented by one or morechips, or can be a random ordering so as to avoid repeating samples ormissing any one sample in the modulo altogether. In fact, themodification of the chip count can be done according to an algorithmprogrammed into the controller or the counter. Such programming would beapparent to a person skilled in the relevant art.

VIII. CONCLUSION

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. It will be apparent to persons skilled inthe relevant art that various changes in form and detail can be madetherein without departing from the spirit and scope of the invention.Thus the present invention should not be limited by any of theabove-described exemplary embodiments, but should be defined only inaccordance with the following claims and their equivalents. All citedpatent documents and publications in the above description areincorporated herein by reference.

1. A method for acquisition and lock of an ultra wideband signal,comprising the steps of: sliding a periodic timing signal using anadjustable time base; producing a decode signal using said periodictiming signal, wherein successive decode signals are coded by successivechips of a code having a predetermined modulo length; cross correlatinga received ultra wideband signal with said decode signal to output abaseband signal; integrating T samples of said baseband signal to outputan integration result, where T is an integer; comparing said integrationresult with a threshold value to output a channel coincidence signal;determining whether channel coincidence has occurred using said channelcoincidence signal, and if channel coincidence is detected, outputting aconstant rate-control signal to stop said periodic timing signal fromsliding, otherwise, if channel coincidence was not detected, adjustingsaid periodic timing signal and repeating the steps of sliding,producing, cross correlating, integrating, comparing and determining forsuccessive T pulses of said periodic timing signal until channelcoincidence is detected.
 2. The method according to claim 1, furtherincluding the step of stopping acquisition if the entire modulo lengthis completed before channel coincidence is detected.
 3. The methodaccording to claim 1, further including the steps of: producing pluraldecode signals; and cross correlating received ultra wideband signalswith said plural decode signals using a plurality of cross correlatorsto reduce the time to acquire channel lock.
 4. The method according toclaim 1, wherein said decode signal is produced by a decode timingmodulator having an address counter, and said adjusting step comprisescounting T pulses of said periodic timing signal and incrementing ordecrementing the address counter by a one or more extra counts aftereach T pulses is reached.
 5. The method according to claim 1, furtherincluding the step of demodulating said baseband signal to output ademodulated information signal.
 6. The method according to claim 5,wherein said demodulating step comprises a step of frequencydemodulating said baseband signal to output a demodulated informationsignal.
 7. The method according to claim 5, wherein said demodulatingstep comprises a step of direct digitally demodulating said basebandsignal to output a demodulated information signal.
 8. A receiver foracquisition and lock of an ultra wideband signal, comprising: anadjustable time base to output a sliding periodic timing signal havingan adjustable repetition rate; a decode timing modulator to output adecode signal in response to said periodic timing signal; a crosscorrelator to cross correlate the ultra wideband signal with said decodesignal and output a baseband signal; first means for integrating Tsamples of said baseband signal to output an integration result, where Tis an integer; a threshold detector to compare said integration resultwith a threshold value to output a channel coincidence signal; and acontroller to determine whether channel coincidence has occurred usingsaid channel coincidence signal, and if channel coincidence is detected,to output a constant rate-control signal to stop said periodic timingsignal from sliding, otherwise, if channel coincidence was not detected,to adjust said periodic timing signal, wherein said means forintegrating and said threshold detector continue to integrate andthreshold detect successive trials of T pulses of said periodic timingsignal until channel coincidence is detected.
 9. The receiver accordingto claim 8, wherein said decode timing modulator having an addresscounter.
 10. The receiver according to claim 9, further comprising: acounter to count T pulses of said periodic timing signal; and extracount logic to increment or decrement said counter by a one or moreextra counts after each T pulses is reached.
 11. The receiver accordingto claim 8, wherein said controller includes means for stoppingacquisition if an entire modulo length is completed before channelcoincidence is detected.
 12. The receiver according to claim 8, furthercomprising: means for producing plural decode signals; and a pluralityof cross correlators to cross correlate received ultra wideband signalswith said plural decode signals to reduce the time to acquire channellock.
 13. The receiver according to claim 8, further comprising ademodulator to demodulate said baseband signal to output a demodulatedinformation signal.
 14. The receiver according to claim 13, wherein saiddemodulator frequency demodulates said baseband signal to output ademodulated information signal.
 15. The receiver according to claim 13,wherein said demodulator direct digitally demodulates said basebandsignal to output a demodulated information signal.